Externally programmable integrated bus terminator for optimizing system bus performance

ABSTRACT

An I/O bus interface cell includes a driver circuit having an input terminal fed by a logic signal and an output terminal to produce in response thereto a drive signal having selectable rise and fall time characteristics in accordance with a reference voltage provided to the driver. The I/O cell also includes a receiver circuit having an input terminal coupled to said output terminal of said driver with the receiver disposed to latch an unresolved, unamplified received signal prior to resolving the state of the signal. The I/O cell further includes a termination circuit having a terminal connected to the output of said driver, and having a selectable impedance characteristic at said terminal, with said selectable impedance being in accordance with a reference voltage provided to an input of said termination circuit. Preferably, the I/O cell the driver, receiver and termination circuits are fabricated on a common semiconductor substrate.

BACKGROUND OF THE INVENTION

This invention relates generally to computer systems and moreparticularly to devices used to drive signals onto and receive signalsfrom a computer bus.

As it is known in the art, computer systems generally include a devicereferred to as a central processor unit which is used to executecomputer instructions to perform some function. The central processingunit generally referred to as a CPU communicates with other devices inthe computer system via a communications network generally referred toas a computer bus or system bus. Other devices commonly coupled to thesystem bus include memory systems such as main memory and morepersistent type of storage systems such as magnetic disk type storagesystems. These devices, including the CPU, are generally not connecteddirectly to the system bus but rather are coupled to the bus through adevice called a bus interface.

The bus interface device for a CPU may be quite different than that fora main memory or for a magnetic disk device. Moreover, for persistentstorage such as magnetic disk, an interface module called a I/O busadapter is often used to interface the system bus to an I/O bus(input/output bus) to which are connected several disk storage devices.In general however, all of these interfaces on a particular bus use acommon set of devices called bus drivers and bus receivers to send andreceive logic signals with proper voltage levels and appropriate drivecapacity to insure reliable transfers of data on the bus.

As it is also known, system buses generally carry information includingaddress information, control information, and data. Buses transfer thisinformation in a logical manner as determined by the design of thesystem. This logical manner is referred to as the bus protocol.

One problem that is common with system buses is that as the performanceof a CPU increases that is, as the processing speed increases, it isnecessary to provide a concomitant increase in bus transfer rate. Thatis, it is necessary to permit more address, control, and data to betransfered at faster rates on the bus so as not to obviate theadvantages obtained by use of a faster CPU.

Buses can be so-called synchronous buses in which all transfers aresynchronized to a common timing signal referred to as a clock signal orthe buses can be asynchronous buses in which hand-shaking signals areused to transfer information as quickly as possible.

Several problems are associated with improving bus performance whetherthe bus is synchronous or asynchronous. A characteristic called cycletime gives an indication of the speed of a bus. For a synchronous bus, acycle can be viewed as that period of time required to complete atransfer on the bus before a new transfer can begin. This minimum perioddetermines the maximum clock rate.

In general, the minimum cycle time for a synchronous bus is related tonoise in the clock generally referred to as clock skew, propagationdelay from an asserting edge of the clock to the period of time that thedata appears at the output of the device connected to the bus, and delayassociated with driving the bus. The delay associated with driving thebus includes two components. The first one is the propagation delaythrough the bus driver and the second is the period of time necessary tohave the bus settle.

Settling time is related to the amount of time necessary to have currenton the bus from the last state of the bus settle out so that the driverfor the next state of the bus can drive current on the bus in theappropiate direction. For example, for a "logic one" level, the driverconnected to the bus typically sinks current from other devicesconnected to the bus, whereas for a "logic zero" the driver typicallsources current onto the bus to other devices connected to the bus.

Bus settling time is also related to the amount of time necessary tohave reflections on the bus die out before data are received byreceivers.

For an asynchronous bus similar electrical considerations are present todetermine a minimum cycle time. For the asynchronous bus, the minimumcycle time for is related to noise in the handshaking signals ,propagation delay from an asserting edge of the handshaking signals tothe period of time that the data appears at the output of the deviceconnected to the bus, and delay associated with driving the bus asindicated above.

Each of the above mentioned electrical parameters are further influencedby tolerances associated with the semiconductor fabrication processesused to provide the circuits interfacing to and feeding signals to thebus. For semiconductor processing certain characteristices such asresistances, capacitances, operating currents, voltages, and propagationdelays, can vary greatly within a lot of devices and over differentperiods of manufacture of the devices. For lower speed buses thisvariation is often not a problem whereas, for higher speed buses, onesolution has been to increase the bus cycle time to take intoconsideration expected processing variations as well as aging effects.The chief drawback with this approach is that it reduces the overallperformance of the system and moreover, it may not be sufficient tohandle all potential variations in characteristices provided bysemiconductor processing.

As data rates on the system bus increases, it has become necessary toterminate the bus in an impedence which mininizes reflections on thebus. Common schemes for terminating a bus require a precise impedencedetermining circuit coupled at each of the two ends of each line of thebus. Termination of a bus in this manner presents a matched impedenceline to signals on the bus. The matched impedance line has the effect ofminimizing reflections on the bus since to the signal the line appearsto be infinite in length. With reduced reflections caused by mismatchedimpedances, less time is required to wait until the signals on the bussettle before data becomes valid and thus speed at which data can betransfered increases.

One problem with this approach to terminating the bus is that by placingthe termination at the end of the bus requires the selection of a highprecision component but does not take easily into considerationvariations between component impedances due to processing variations orchanges over time. Further, termination of a bus at the ends of the busstill leaves portions of the bus between common bus conductors andindividual devices on the bus unterminated. This can become a source ofreflections on the bus as speeds on the bus increaes. These reflectionsare not adequately compensated for by the above mentioned approach.

Therefore, while use of a precise terminating impedance such as anexternally coupled resistor generally provides a suitable impedance atlower operating frequencies of a bus, such an external termination stillcan require a system designer to allocate sufficient time for the bus tosettle for buses operating at higher frequencies. Moreover, thistechnique is also suceptible to providing less than optimum performanceat high frequencies due to variations in device characteristics causedby processing variations from device to device.

Further, such a technique does not easily allow for changing thetermination resistance of a complicated bus. For example, a modificationof a bus design may require a different resistance to terminate the bus.That is often bus termination resistance is determined on prototypedevices. When actual devices are built however, the optiminaltermination resistance may be different. To change the terminationresistance requires a redesign of the bus interface devices andfabrication of a new lot of such devices. This is expensive and timeconsuming and hence undesirable.

SUMMARY OF THE INVENTION

In accordance with the present invention, a bus interface circuitincludes a plurality of I/O cells, each cell including terminationmeans, responsive to a voltage reference signal representative of adesired impedance value, for providing the desired termination impedancefor the cell. The bus interface circuit further includes means,responsive to a resistance, for providing said voltage reference signalto said termination means of each of said I/O cells. With such anarrangement, a resistance is used to provide a reference voltage to setresistance values of each termination means in each cell of an I/Ocircuit. Thus, by changing the value of the resistance, the value of thereference voltage and hence the value of the termination resistance ofeach cell is also changed. This makes changes to a bus network easier toaccomplish with out a redesign or refabrication of the network.

In accordance with a further aspect of the present invention, the meansfor providing the voltage reference signal of the circuit includessecond termination means, substantially identical to said terminationmeans in each of said I/O cells, responsive to a second voltagereference signal corresponding to a known system voltage referencesupply, for providing the voltage reference signal by providing atermination impedance at the output of said second means correspondingto said resistance. With such an arrangement a technique is alsoprovided to calibrate the reference signal to a termination means whichis substantially identical to that used in each of the I/O cells. Thisprovides a reference voltage which is thus also representative of adesired impedance to be provided at the output of each of thetermination means of the plurality of cells.

In a still preferred embodiment the circuit is fabricated as anintegrated circuit and the resistance is a single resistor disposedexternal to the integrated circuit. With such an arrangement, theexternal resistance can be changed as necessary to provide a newtermination resistance for each of the cells without making substantialchanges to the circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

The above-mentioned and other features of the invention will now becomemore apparent by reference to the following description taken inconnection with the accompanying drawings in which:

FIG. 1 is a block diagram of a computer system including a system bus;

FIG. 2 is a diagrammatical representational view of a layout of portionsof integrated circuit used in the computer system of FIG. 1, showing businterface circuit portions of the integrated circuits coupled to thesystem bus of the computer system of FIG. 1;

FIG. 3 is a block diagram of an integrated bus I/O cell includingdriver, receiver, and termination circuits to interface to the systembus shown in FIGS. 1 and 2;

FIG. 4 is a block diagram of a termination circuit used in theintegrated bus I/O cell of FIG. 3;

FIG. 5 is a diagram showing the relationship between FIGS. 5A and 5B;

FIGS. 5A and 5B are schematic diagrams of an embodiment of thetermination circuit of FIG. 4;

FIG. 6 is a block diagram of a control circuit used to provide areference voltage for the termination circuit and driver circuit of FIG.3;

FIG. 7 is a block diagram of a termination reference voltage controlcircuit used to provide a reference voltage for the termination circuitof FIGS. 4 and 5;

FIG. 8 is a block diagram of a driver used in the integrated bus I/Ocell of FIG. 3;

FIG. 9 is a diagram showing the relationship between FIGS. 9A and 9B;

FIGS. 9A and 9B are schematic diagrams of an embodiment of the driver ofFIG. 8;

FIG. 9C is a block diagram of a control circuit used to provide a signalto control current from the driver of FIGS. 9A and 9B;

FIG. 10 is a schematic diagram of a clamping circuit for the driver ofFIG. 8;

FIG. 11 is a schematic diagram of a speed cell circuit for the driver ofFIG. 8;

FIG. 12 is a timing diagram useful in understanding the speed cellcircuit of FIG. 11;

FIG. 13 is a block diagram of a logic state device used in the driver ofFIG. 8;

FIG. 14A is a block diagram of a bus receiver;

FIG. 14B is a block diagram of an embodiment of the receiver of FIG.14A;

FIGS. 15A and 15B are timing diagrams useful in understanding signaltiming relationships in the receiver of FIG. 14B;

FIG. 16 is a block diagram of a data resolution circuit used in thereceiver of FIG. 14B;

FIG. 17 is an electrical schematic of a resolver circuit used in thedata resolution circuit of FIG. 16;

FIG. 18 is an electrical schematic diagram of an embodiment of thereceiver of FIG. 16;

FIG. 19 is a block diagram of a primary-replica amplifier having lowoffset voltage;

FIG. 20 is a schematic diagram of an operational amplifier used in theamplifier of FIG. 19;

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring now to FIG. 1, a computer system 10 is shown to include a pairof central processing units (CPU's) 12, 14 each CPU including aninterface circuit 12a, 14a respectively. The computer system 10 is shownto further include here four memory modules 16, 18, 20 and 22 each alsohaving interface circuits 16a through 22a respectively and three I/Omodules 24, 26, and 28 with each having respective interface circuits24a, 26a and 28a, as shown. Each of the interfaces 12a through 28a areinterconnected together via a system bus 30. System bus 30 includesaddress, control and data lines as necessary. The interfaces for each ofthe devices 12 to 28 include logic circuits (not shown) to provideproper logic processing functions of logic signals provided via the bus30, as appropriate according to a designated protocol implemented by thebus 30. These logic circuits (not shown) are coupled between circuits onthe respective modules and bus interface circuits, as will be described.

Referring now to FIG. 2, a representative portion of the system 10 isshown. Here modules 14 and 24 (FIG. 1) are shown diagrammatically asintegrated circuits 14' and 24' each having a plurality of integratedbus I/O interface cells 40 disposed about the periphery of theintegrated circuits. Details of the interface cells 40 will be discussedin conjunction with FIGS. 3 to 20.

Suffice it to say that the devices 14 and 24, of FIG. 1 include anintegrated circuit 14a' and 24a', here an ASIC (application specificintegrated circuit) which is part of the interface circuit 14a and 24ato the particular device 14, 24 (FIG. 1). Each I/O cell 40 of eachcircuit for each device is interconnected via conductors of bus 30 whichare typically disposed on a printed wiring board (PWB) such as abackplane or motherboard (not shown) to each corresponding line on thedevices 12 to 28 (FIG. 1). Each of the bus interface cells are alsointerconnect via a interface cell control bus 41, 41' with each of saidbuses typically being unique for each one of said devices 12-28. Each ofsuch buses include reference voltage signals, enable signals and supplyvoltage signals, as necessary for operation of integrated I/O cellcircuits, as will now be described.

Referring now to FIG. 3, an illustrative one of the bus interface cells40 is shown to include a programmable internal termination resistance42, a driver 70 and a receiver 90. The programmable internal terminationcircuit 42 has an IO output terminal 43 which is connected to respectiveIO terminals on the driver 70 and the receiver 90 at a point proximatethe latter two devices. Since the termination circuit 42 is preferablyintegrated on the same semiconductor circuit as the driver 70 and thereceiver 90, the distance between the connections is extremely small.

The bus interface cell 40 is fed by a first plurality of lines 40a to40g corresponding to voltage reference signals, control signals andvoltage potentials which are common to all of the cells 40 on a givenASIC.

Signal "VOH" on line 40a is "voltage output high" and is used by thedriver 70 and an overshoot clamp overshoot 86 circuit in the termination42 to specify the maximum output high voltage for a logic "one" state.That is, this signal is a voltage reference corresponding to a limit ofthe driver high level voltage in an unloaded state. Signal "VDDXTL" online 40b is the external supply voltage used by the termination circuit42, the driver circuit 70, and receiver circuit 90 as the basic supplyvoltage for the circuits. In some implementations of the integrated IOcell it is preferable to have a separate supply connection for thereceiver 90. Signal "VDRVCTL" on line 40c is "voltage driver controlsignal" and is used to determine the level of current driven by thedriver. Signal "VDACTL" on line 40d is voltage reference for adifferential amplifier in the driver 70 and is used by the driver 70 tocontrol the differential amplifier current. In particular, signal VDACTLis used to control the rise and fall times of a signal driven from thedriver device 70. Signal "VTERMCTL" is the voltage termination controlsignal and is fed on line 40e to the termination circuit 42. SignalVTERMCTL is used to represent the desired resistance characteristic tobe provided by the termination circuit 42. This voltage signal isconverted into a resistance within circuit 42 to provide in combinationwith a fixed resistance (not shown) in circuit 42 the desired resistancecharacteristic. Signal "RCVREF" on line 40f is the receiver referencelevel and is used by the receiver 60 to distinguish between a logic highlevel and a logic low level. Signal "REFSINK" on line 40g is thereference sink level and is used by the receiver 90 during prechargingof a data resolution circuit in the receiver.

Signals VTERMCTL and VOH and VDDXTL are used by the termination circuit42 to provide a controlled, compensated termination resistance at thepad IO. This permits each line of bus 30 to be terminated in acharacteristic impedance which minimizes reflections on the bus 30 andalso minimizes reflections on lines between the system bus 30 and thedriver/receiver circuits in each of the cells 40. Thus, the length ofconductor between the driver and, receiver circuits 70, 90 and the bus30 is also properly terminated. Moreover, by providing a terminationcircuit 42 which provides a programmable impedance, here a resistancetype, the circuit can compensate for variations in circuit parameterscaused by differences in circuit processing, aging and temperature.

The signal VTERMCTL is derived from an externally settable resistance52, RTERM, (FIG. 2) which is used to calibrate the signal under changingconditions. Derivation of the signal VTERMCTL will be discussed inconjunction with FIG. 6 and 7.

Driver 70 here is a voltage mode controlled device. The driver 70 isused to drive a signal on to the bus 30. The driver 70 operates from astandard (non-reduced) supply voltage to provide I/O signals withreduced voltage swings. The driver 70 further provides the driven signalas a logic "1" (one) level. Thus the driver sources current when drivingthe bus to a logic "1" level. The termination 42 is used to pull the busline down to VSS to provide the logic "0" state. The driver furtherprovides signals having controlled rise and fall times to furtherincrease bus speed. The rise and fall times are controlled via a singleexternal resistance 53 (RRISE) for each module. Details of the circuitsused to provide external control of the rise and fall times of thedriver 70 will be discussed in conjunction with FIG. 9C. The driver 70includes a speed increase circuit to anticipates the next state of thebus. Thus, the speed increase circuit preconditions the bus to increasebus speed.

Receiver 90 is a precision receiver which latches an unamplified,unresolved state of a signal at an input thereof.

Referring now to FIG. 4, the termination circuit 42 is shown to includea voltage to resistance translation circuit 44 which is fed by signalsVTERMCTL, TENB, and SDRV. The termination circuit 40 also includes again reduction circuit 46 and an active resistance circuit 47. Theactive resistance circuit includes a transistor 47g coupled in serieswith a fixed, passive resistance 49 and coupled in shunt with theremainder of the active resistance circuit 47. The transistor 47gcoupled in series with the fixed passive resistance network 49 providesa resistance characteristic which is variable between the value of thefixed passive resistance and maximum resistance of the transistor 47g,and the minimum resistance provided by the combination of the fixedpassive resistance and the resistance provided by transistor 47g and theresistance of active resistance circuit 47 in accordance with controlsignals provided from voltage to resistance translation circuit 44.

A linearizing circuit 48 is disposed in parallel with the transistor 47gto improve the linearity of the transistor 47g and active resistancecircuit 47.

Voltage to resistance translation circuit 44 converts the signalVTERMCTL to a series of output signals on lines 44a-44h. These signalson lines 44a-44h are here representative of threshold voltage levelswhich when applied to active resistance circuit 47 will produce inresponse, selectable levels of resistance at the output of the circuit47. The selectable levels of resistance in parallel with the transistor47g provide a selectable resistance characteristic in series with thefixed passive resistance 49.

Signals 44a-44h are preferably fed to a gain reduction circuit 46. Gainreduction circuit 46 provides a resistance connecting each of theadjacent control signals. Since the voltage to resistance translationcircuit 44 provides an output voltage which does not vary linearly withapplied input voltage as provided from the signal VTERMCTL, the gainreduction circuit 46 introduces a device based resistance into the pathbetween the voltage to resistance translation circuit and the activeresistance circuit. This makes the relationship between VTERMCTL andoutput resistance from the active resistance more linear.

The output of the translation circuit 44 or the optional gain reductioncircuit 46 is fed to the active resistance 47 to provide, in combinationwith the fixed resistance 49, a selectable resistance characteristic atthe output of the termination circuit 40 at terminal 30a (IO), asmentioned. Here, fixed passive resistance 49 is comprised of a firstresistance 49a and a second resistance 49b. The second resistance isdisposed between terminal IO and a terminal IORES as shown.

Referring now to FIGS. 5A and 5B an exemplary one of the terminationcircuits 42 is shown to include the voltage to resistance translationcircuit 44 which includes a plurality of successively coupled stages ofcircuits here inverters 44a to 44h. Each inverter includes a pair oftransistors here complementary metal oxide semiconductor (CMOS)transistors. That is, each inverter 44a to 44h includes an NMOS deviceand a PMOS device with NMOS devices denoted by a direct line between agate and drain-source junction and each PMOS device denoted by an opencircle in the gate. Further each of said transistors have an indicatedsize, with W corresponding to a width of the device and L correspondingto a length of the device. Further the size is expressed inComplementary Device Units where one unit corresponds to 0.375 microns.Each of the transistors includes a control electrode or gate electrodeand a pair of output electrodes or source and drain electrodes. Each ofthe gate electrodes is connected to the line carrying signal VTERMCTL;whereas the source electrodes of the NMOS devices are here connected toa reference potential "VSS" and the drain electrodes of the NMOS devicesare connected to control signal terminals T1 to T8 respectively asshown. The drain electrodes are also connected to the drain electrodesof a corresponding one of the PMOS transistors as shown. The sourceelectrodes of the PMOS devices are connected to line 40f carrying signalTENB. The transistors of the inverters also have a connection to thesubstrate or bulk material of the PMOS devices due to processconsiderations of PMOS devices. This connection (not shown) is made toVSS the supply voltage for NMOS devices.

Signal TENB is provided from the receiver (FIG. 13) as will bedescribed. TENB is used to switch off the termination when the lineassociated with the termination is being driven. Thus, for the abovearrangement since the termination is coupled to VSS or as here groundpotential, the driver drives current onto a corresponding line when thestate of the signal is a logic one. Thus, TENB is used to turn off thetermination such that the driver 70 (FIG. 3) does not drive current intoits own termination resistance provided by termination circuit 42 whenthat driver is placing a logic one onto the output line IO.

When TENB is in a high state and SDRV is low, this state indicates thatthe driver is not driving the line IO and the sources of transistors P13to P20 are at a high voltage level and thus the converter is on. WhenTENB is low and SDRV is high this indicates that the driver is drivingthe line IO and thus turning the termination off. This arrangement savespower while driving since the driving cell 40 does not drive currentinto its own termination. Further this also reduces power dissipation inthe driver circuit which is driving the bus line and concomitanttherewith reduces dissipated heat.

Each of the inverters 44a to 44h is thus successively connected betweenTENB and VSS. The inverters 44a to 44h each have selected sizes toprovide concomitant values of threshold voltage to turn on the NMOStransistors 47a-47f at different voltage levels. The selected values ofthreshold voltage are provided by scaling the ratios of the sizes of thePMOS transistor to the size of the corresponding NMOS transistor(expressed in CMOS drawn units or CDU's). A CDU is the equivalent of0.375 microns. Here the inverters are arranged in successivelyincreasing threshold characteristics or successively increasing ratiocharacteristics. An illustrative example of values of ratios is setforth in Table I below.

                  TABLE I                                                         ______________________________________                                                        PMOS IN      NMOS in                                          INVERTER        CSU's        CDU's    P(W/L)/                                 STAGE      W        L     W       L   N(W/L)                                  ______________________________________                                        44a         8       8     16      2   1/8                                     44b         8       4     12      2   1/3                                     44c        12       2     16      2   3/4                                     44d        12       2     8       2   1.5                                     44e        13       2     8       4   3.25                                    44f        22       2     8       4   5.5                                     44g        48       2     8       4   12                                      44h        64       2     8       8   32                                      ______________________________________                                    

The voltage to resistance translator 44 further includes a thirdplurality of here NMOS transistors coupled in shunt with the second andsucceeding ones of said first plurality of NMOS transistors. Since thesecond plurality of NMOS transistors are relatively small in size orhave a relative low gate drive, the third plurality of 44 transistorsare used aid the corresponding NMOS transistors of the inverters 44b to44h to,turn off or pull lines T1 to T8 to VSS.

In operation circuit 44, a voltage signal VTERMCTL on line 40e is fed tothe gate electrodes of each of the transistors in the circuit 44. Inaccordance with the value of the signal VTERMCTL, possibly the firststage and here consecutive succeeding ones of the stages will turn "on"or conduct with the NMOS, PMOS transistors of the respective stageproviding a voltage at the output of the respective stage and thus onrespective lines T1 to T8. Succeeding ones of the inverters will havethe PMOS transistors turn on until the value of the VTERMCTL voltage isless than the threshold voltage of the inverter, causing the PMOS deviceto turn on and place a logic one on the succeeding output lines throughT8. In this manner a voltage level representative of a desiredresistance value is translated into a series voltage level controlsignals.

These voltage level control signals are fed to the gain reductioncircuit 46. The gain reduction circuit 46 includes a second plurality ofstages. Each of the stages includes a pair of coupled PMOS and NMOStransistors with corresponding gate electrodes coupled to VSS or VDDrespectively. The gain reduction circuit is fed by signals T1 to T8 asshown. A signal on line T2 is fed to the first pair of transistors whichbeing in a resistive mode adds a transistor based resistance into thepath provided from the voltage to resistance translation circuit 44 tothe active resistance circuit 47. This arrangement is used to flattenout the response of the first stage of the voltage to resistancetranslation circuit 44 reducing the gain from the stage and causing achange in resistance between successive lines to be smaller.

The second and succeeding signals except the last signal, are alsocoupled to the active resistance circuit 47. Active resistance circuit47 includes a plurality of NMOS transistors and each having selectedsizes. A illustrative example of sizes is given in Table II.

                  TABLE II                                                        ______________________________________                                        DEVICE  WIDTH (CDU)  LENGTH (CDU) RATIO W/L                                   ______________________________________                                        47a     86           2            43                                          47b     78           2            36                                          47c     55           2            27.5                                        47d     54           3            18                                          47e     24           3            8                                           47f     18           4            4.5                                         ______________________________________                                    

The control signals T2 to T7 are fed to the control electrodes (gates)of each of the transistors 47a-47f respectively. The respective drainsand sources of the transistors 47a to 47f are coupled together, asshown, to provide a network of parallel connected devices. The activeresistance 47 further includes an NMOS transistor g which is coupled inparallel with the remaining transistors 47a to 47f as shown. The gate oftransistor 47g is fed signal TENB.

Transistor 47g is selected to have a size such that when TENB isasserted "high" transistor 47g provides the termination circuit 42 witha maximum active resistance of 83 ohms. This maximum resistance incombination with the resistance of the fixed resistance 49 provides anoverall maximum resistance of about 180 ohms, here used at extremevariations caused by semiconductor processing, supply voltage andoperating temperature. Successive ones of the remaining transistors arecoupled in parallel with the transistor 47g to selectively reduce theresistance across the nodes to which transistor 47g is connected to aselected value in accordance with the state of the control signalsderived from the reference voltage signal VTERMCTL. Transistor 47g isalso coupled to fixed resistance network 49 and VSS. Fixed resistance 49provides a fixed passive resistance of about 97 ohms although othervalues could be used.

The combination of fixed resistance 49 and active resistance 47 providesthe termination with a resistance which can vary between a minimum asdetermined by the minimum resistance of network 47 (here 29 ohms) andfixed resistance 49 (here 97) ohms and a maximum as determined by themaximum resistance of network 47 (here 83 ohms) and fixed resistance (49here 131 ohms).

As shown in FIG. 5B, a termination switching circuit 43 is also disposedaround transistor 47g. Circuit 43 includes two transistors 43a and 43b.Transistor 43a has its gate electrode coupled to TENB whereas transistor43b has its gate electrode coupled to signal SDRV. Transistor 43acontrolled by signal TENB and transistor 43b controlled by signal SDRVboth act as switches. As mentioned above, signal TENB is also fed totransistor 47g. When TENB is asserted high and SDRV is low the activeand fixed resistances are connected to the output of the circuit. Thispermits the termination to terminate the signal provided to theparticular I/O cell.

When SDRV is high and TENB is low, the termination is removed from thecircuit which is driving its corresponding I/O line 30a to a logic "1"level. This reduces power dissipation and hence heat in those I/O cellswhich are driving the bus 30.

The transistors 43a and 43b have source and drain electrodesrespectively tied together and coupled to a gate electrode of atransistor 48. Thus, the state of these transistors is used to controlthe gate of transistor 48. The transistor 48 has its gate drivenessentially from the output signal at terminal IORES, ie. throughtransistor 43a. The drain of transistor 48, however, is connected to theconnection of the active and passive resistance networks that is at avoltage divider. Thus the operating characteristics of transistor 48 aredifferent than those of transistor 47g and transistors 47a to 47f inactive resistance network 47. This permits the transistor to compensatefor the non-linear characteristics of the active resistance network 47.

Transistor 48 has a resistance versus voltage characteristic which isopposite to that of the transistors in the active resistance network 47.Thus, when transistor 48 is enabled, it provides a non-linear resistancecharacteristic which tends to be opposite to and thus compensates forthe non-linear resistance characteristic of the active resistancenetwork 47. This arrangement presents a more linear resistancecharacteristic to a signal fed to the terminal IO. The linerairycharacteristic of transistors 47 is such that as voltage at signal IO isincreased, resistance of transistor 47a to 47g increase. Opposing thisthe linearity characteristic of transistor 48 is such that as voltage atsignal IO increases the resistance of the transistor 48 decreases.Transistors 47a-47g and 48 are coupled at source and drain terminals toprovide in combination a characteristic resistance which is more linearvs. applied voltage.

Referring now to FIG.6, a control circuit 50 to control the setting ofthe reference voltage VTERMCTL is shown to include a preferably highprecision, highly stable resistance 52 connected external to the controlcircuit 50 and to its associated ASIC 12a-28a (FIG. 1). The resistance52 has a terminal coupled to a non-inverting input of an operationalamplifier 54, here a compensated amplifier having a low offset voltageand a remaining terminal coupled to VSS. The inverting input of theamplifier 54 is coupled to a terminal which is fed by a precise andstable reference voltage, VIH. Voltage VIH is the reference voltagecorresponding to a representative high input voltage level. The circuit50 is shown to further include a transistor 56 having a gate electrodecoupled to the output of the amplifier 54 and a source electrode coupledto VDDXTL, the external supply voltage source. The transistor 56 isshown to further include a drain electrode coupled to the externalresistance 52, as shown.

The above arrangement is used to drive the output of the amplifier 54 toa voltage "VDRVCTL" which corresponds to that voltage required to insurethat the voltage at the two inputs of the amplifier 54 are equal. Thus,given a reference voltage VIH, the amplifier output via the connectionto the gate electrode of transistor 56, drives the transistor 56 to avoltage at the drain across external resistance RTERM that equals VIH.As process, aging, and operating conditions vary, the voltage on theoutput of the amplifier also varies to provide VDRVCTL tracking allsources of variation.

voltage VDRVCTL is used to control the operating current of speedincrease circuit 82, (FIG. 8), the driver 70 current limit, as well asto provide timing control for the circuit which provides VDACTL (FIG.9C). The current controlled by VDRVCTL is unvarying for a fixed VIHvoltage and RTERM resistance, and is identical in all ASIC's. Thiscontrolled current is proportional to voltage VIH divided by resistanceRTERM. The driver current limit is preferably eight (8) times thecurrent flowing through RTERM, representative of a driving modulesupplying driver current to eight receiving, and hence terminatingmodules each disposed at an RTERM terminating, resistance.

Referring now to FIG. 7, the reference voltage VIH₋₋ PROT is also fed toa non-inverting input of a second operational amplifier 57, here also acompensated amplifier having a low offset voltage. An inverting input ofthe amplifier 57 is fed from a drain electrode of a transistor 58.Transistor 58 further has a gate electrode fed by the signal VDRVCTLfrom operational amplifier 54 (FIG. 6) and a source electrode connectedto the external supply voltage VDDXTL. The output of the amplifier 57 iscoupled to reference input terminal "VTERMCTL" of a termination circuit59 which is disposed as part of the control circuit 50 of the businterface for the device. Since the control circuit is preferably on thesame integrated circuit as the I/O cells 40 for the interface for thecircuit, the termination 59 in the control circuit will havesubstantially the same characteristics as the termination 42 in each ofthe I/O cells 40.

The signal VDRVCTL from amplifier 54 drives transistor 58 which ispreferably the same size as and has the same characteristics astransistor 56 to produce a voltage on the inverting input to theamplifier 57. The amplifier 57 drives the line VTERMCTL through thetermination 59 so that the value of the signal VTERMCTL corresponds tothat necessary to set the resistance of the termination 59 in controlcircuit 50 to the same resistance as RTERM (FIGS. 2 and 6). The voltagereference VTERMCTL thus provides a calibrated signal which is used toset each termination 42 in each I/O cell 40 of a particular ASIC tosubstantially the same value of resistance.

Thus, with this arrangement, only a single resistance RTERM per ASIC isneeded to provide a common reference voltage for the particular ASICwhen setting termination resistance. The signal VTERMCTL is derived froma stable fixed resistance, precise reference voltage, and arepresentative termination circuit 59, which, being on the same chip asthe terminations 42 in each I/O cell provides the termination 59 andtermination 42 with substantially the same electrical characteristic.Accordingly, this arrangement can be used to compensate for electricalvariations caused by aging, temperature, supply voltage and processvariations.

Therefore, the value of VTERMCTL is set with respect to a preciseresistance which is provided external to the termination 42 and thevoltage value of VTERMCTL is also calibrated against a representativeone of the termination circuits 59 to insure that the voltage referencesignal VTERMCTL reproduces a similar resistance on the output of each ofthe terminations 42 in the I/O cells 40.

Referring now to FIG. 8, a driver 70 having controlled rise and falltime characteristics is shown to include a logic state device 72, here aconventional master slave D-type flip flop. Logic state device 72 is fedby a clock signal CLK, an enable signal EN, and a data signal D. Thesesignals are logic-type signals provided from a logic circuit (not shown)which the driver 70 interfaces to in one of the modules 12-28 of FIG. 1.From the output of the logic state device 72 is provided a pair ofterminals SDRV and SDRV₋₋ L. Here SDRV is fed to a turn-on chargingcurrent control circuit 74 and a turn-off charging current controlcircuit 76. The turn-on charging current control circuit 74 and turn-offcontrol charging current circuit 76 are also fed via reference voltages"VDACTL" (FIG. 9C) "VOH" (FIG. 9C) and a electro-static protectedvoltage signal "IOPROT". IOPROT is provided from signal IO via aresistance on the receiver 90 (FIG. 14).

When a data signal is fed to logic state device 72, the state of thesignal is clocked into the device via the clock signal along line CLK.Alternatively, the signal along line CLK could be an asychronoushandshaking signal developed by an asychronous interface (not shown). Ata predetermined period of time after the asserting edge of the clocksignal, the data appears at the outputs SDRV and SDRV₋₋ L of the logicstate device 72.

Turn-on control circuit 74 is used to control the rise time of theoutput signal of a driver device 80, whereas, turn-off control circuit76 is used to control the fall time of the driver device 80.

The turn-on control circuit 74 in response to reference signals VOH,IOPROT, VDACTL and data signals SDRV₋₋ L from logic state device 72provides an output signal along line 80a to charge an input control lineto the driver device 80 at a current proportional to the input voltagerequired to develop a constant reference output current from driverdevice 80. This provides a fixed time period over which to charge thedriver input, thus providing a controlled output rise time from thedriver.

Correspondingly, the turn-off control circuit 76 is fed a data signalvia the SDRV₋₋ L output from logic state device 72, as well as referencesignals VDACTL and VOH. The turn-off control circuit 76 likewiseprovides a signal used to discharge the input line of the driver deviceat a current proportional to the driver input voltage required to returnthe output of the driver from the driven voltage to a reference voltageor "off state" over a fixed period of time. This also results in a fixedperiod to discharge the driver gate, thus controlling output fall timefrom the driver device 80 and hence the driver circuit 70.

Charge current turn on control and turn-off control circuits 74 and 76are also coupled along lines 74a and 76a to a disable circuit 81 and avoltage source mode control circuit 73, respectively, as shown. Thedisable circuit 81 is used to disable the charging current turn oncircuit 74 after a predetermined period of time as will be furtherdescribed below.

The driver 70 further includes, in addition to the voltage source modecontrol circuit 73, a current source mode control circuit 75, as well asvoltage limit control circuit 77 and current limit control circuit 78.The voltage source mode control circuit 73, V-limit control circuit 77,current source mode control circuit 75, and I-limit control circuit 78are each coupled to the input of the driver device 80.

Each of the aforementioned circuits is used to provide selective controlto the input or gate electrode of the driver device 80. The driver 70thus has circuits to permit the driver 70 to operate alternatinglybetween a voltage source mode and a current source mode when driving asignal from the output 30a of the driver device 80.

Here a relatively large PMOS transistor is used as the driver transistorof driver device 80, although other types of transistors couldalternatively be used. Since a PMOS device is used as the driver device,a high voltage relative to VSS is fed to the gate thereof to turn thedevice off whereas, a low voltage relative to VDD is used to turn thedevice on. Further, since the termination for each I/O cell is coupledto VSS, the driver device 80 drives a logic 1 level by sourcing currentto the termination from the VDD supply and releases the line allowingthe termination to pull the line down to VSS for a logic zero level.Thus, when placing a logic one on the line 30a, the driver device 80sources current from VDD.

The driver operates as follows: When driving a logic one from the driverin response to the asserting transition of data from the logic statedevice 72, via line SDRV₋₋ L, turn-on charge control circuit 74 providesa charging current signal on DRVR₋₋ GATE line 80a to start lowering thevoltage therein.

Since the voltage on DRVR₋₋ GATE line 80a is initially higher inmagnitude than VDRVCTL, I-limit control circuit 78 asserts a signalwhich enables the voltage source mode control circuit 73 to also providean output signal on line 80a. Turn-on charge control circuit 74 isturned off either by disable circuit 81 or by monitoring IOPROT when itis above about 1 V. The voltage source mode control circuit 73 lowersthe voltage on DRVR₋₋ GATE line 80a driving driver transistor 80 towardsa logic "1" level. The voltage source mode control circuit 73 thereafteradjusts the signal magnitude on DRVR₋₋ GATE line 80a in accordance withvoltage reference signal VOH and output voltage IOPROT. The circuit 73uses VDACTL to set a current level from the output of the voltage sourcemode device 80. Whereas, IOPROT is used by the voltage source modecontrol circuit 73 to adjust the voltage on hDRVR₋₋ GATE line 80a toprovide a constant output voltage relative to VOH from driver device 80in accordance with loading of the driver device 80.

The V-mode control circuit 73 continues to drive the transistor 80 whilethe magnitude of the voltage on line 80a is also monitored by theI-limit control circuit 78. When the voltage on line 80a approaches oris substantially equal to the voltage of VDRVCTL, the I-limit controlcircuit 78 will inhibit the v-mode control circuit 73 from driving theDRVR₋₋ GATE line 80a and, correspondingly, the I-limit control 77circuit will permit the I-mode control circuit 75 to drive line 80a viasignal OC₋₋ L. This signal is low when driver is at current limit.

When the magnitude of the drive signal on line 80a exceeds a voltagewhich would cause the output of a transistor 80 to exceed VOH, thevoltage limit control circuit 77 causes the output drive line 80a todrop to a voltage below that which would drive transistor 80 to avoltage above VOH by comparing the values of voltages VOH and IOPROT.

I-mode control circuit 75 maintains a signal on line 80a to followchanges in VDRVCTL. The V-limit control circuit 78 compares the voltageon the output of driver device 80 to the magnitude of reference signalVOH. If the voltage on the output IO exceeds VOH, then the V-limitcontrol circuit 77 increases the drive voltage on the DRVR₋₋ GATE line80a to reduce the output voltage from the device. This turns off theI-mode control 75 and turns-on the V-mode control 73 when the I-limitcontrol 77 detects the voltage of DRVR₋₋ GATE being greater than thevoltage of VDRVCTL.

Accordingly, by use of voltage source mode control circuit 73, I sourcemode control circuit 75, V- limit circuit 75 and I-limit control circuit77, the driver transistor 80 is switched between a voltage source modeand a current source mode when driving the driver device 80 to a logicone state, as load current demand varies from below current limit toabove current limit.

The driver releases the line IO to provide a logic zero state when thelogic state device 72 provides a logic 1 on SDRV₋₋ L. The signal is fedto the turn on control circuit 74 and the turn off control circuit 76,as well as the voltage source mode control circuit 73 and the V-limitcontrol circuit 75. These circuits in response cause the signal to thedriver device to go to a high voltage level, thus allowing the driverdevice to release the output line whereas the terminations on other endsof the line (part of the bus interfaces for the other devices) and ofthis IO cell 70 now pull the line to VSS providing a logic zero level.

The driver 70 further includes an overshoot clamping circuit 83 coupledto the output 30a of driver 70. The overshoot clamp circuit 83 limitspositive and negative transient voltage excursions on the output line30a. The clamping circuit 83 limits positive overshoots by diverting bussignal current into the VSS supply node through a transistor (notshown). The clamping circuit 83 limits negative overshoots by decreasingdrive voltage on DRVR₋₋ GATE line 80a to prevent complete turn-off ofthe driver transistor 80 until the current on line 30a falls to nearzero.

The clamping circuit 83 is provided to prevent transients on signallines where multiple drivers can drive the line IO, particularly forpositive overshoots, and to limit negative voltages caused by transientsassociated with inherent inductance characteristics of the conductorcomprising the line 30a (IO).

The driver 70 further includes a speed increase circuit 82 coupled tothe output 30a of the driver circuit 70. The speed circuit 82 is fed alogic signal from an LZI output of the receiver 90 (FIG. 3) to sense thelogic state most recently received at the receiver 90. The speed circuit82 is also fed signals PEDALN and VDRVCTL. In response to these signalsthe speed circuit 82 provides, in a portion of the bus cycle afterreception of the most recently received logic signal an output signal toprecondition the line 30a to the next state of the BUS. Here, the speedcircuit 82 operates such that the next state of the line 30a on the bus30 is the opposite logic state to the state most recently received bythe receiver 90.

By preconditioning the output of the line 30a of BUS 30, less voltageswing is required on the line 30a during a change of state, thusimproving performance by reducing the amount of time necessary to changelogic states on the line 30a. If the next state of the bus is the sameas the previous state, then the preconditioning continues so that whenthe transition occurs it will be at an increased speed. No significantpenalty is incurred since the bus remains in the correct state.

Referring to FIGS. 9A and 9B, an illustrative example of the drivercircuit 70, generally discussed in conjunction with FIG. 8, is shown toinclude the turn-on control circuit 74 including a threshold detectorcircuit 74a which is coupled to a transistor stack 74b. Here eachtransistor is denoted by a reference designation such as P1 which isunique for the particular Figure. Other figures may use the samereference number when designating transistors in the circuit describedin the figure. Further, from here on in each transistor is designated bya size with the first number being the width and the second number,separated by a comma "," being the length.

The threshold detector 74a includes an NMOS transistor P20 coupled to aPMOS transistor P18 which is coupled to a PMOS transistor P17. PMOStransistor 17 has a gate electrode which is fed by the signal DISABLEfrom the disable circuit 81 whereas transistors P18 and P20 have gateelectrodes coupled to SDRV₋₋ L. The threshold detector further includestransistors P19 and P21 which are used in turning off the circuit.

The stack 74b includes transistors P22 through P24, as shown. The gateof transistor P22 is coupled to the drain of transistor P19, whereasgates of transistors P23 and P24 are coupled to respectively VOH andVDACTL. The relative values of the voltages VOH and VDACTL determine thecurrent flow through the stack when the transistor P22 is turned on.

A circuit used to set the value of VDACTL and hence the rise and falltime of the driver and the output current from the driver will bediscussed in conjunction with FIG. 9C.

Initially, assuming that SDRV₋₋ L is high then transistor P17 is on(since the disable circuit via line disable, has not yet disabled theturn-on charge control circuit) and transistor P18 is off. As SDRV₋₋ Ltransitions low, transistor P18 is driven on while transistor P20 isdriven off. At this juncture, transistor P19 is still off since its gateelectrode is coupled to IOPROT which is still in a low state due to theprevious state on the bus as determined from SDRV₋₋ L having been at alogic one and transistor P21 is also off since the disable output ishigh.

In response to the signal SDRV₋₋ L turning on transistor P18, transistorP18 causes a large current to be drawn from transistor P22 via a signalon the gate electrode of transistor P22. Transistor P22 is sized tosource a large current. Hence, transistors P18 and P22 sink a currentfrom DRVR₋₋ GATE line 80a. The amount of current on DRVR₋₋ GATE line 80ais determined in accordance with the values of voltage of VOH and VDACTLfed to the gate electrodes of transistors P23 and P24.

Transistors P19 and P20 are used to control and hold turn-off of thecharge control circuit 74. Transistor P21 turns off the charge controlcircuit 74 via a signal from the disable circuit 81.

Disable circuit 81 includes here two inverters (transistors P46 and P47,and transistors P43 and P44) coupled together, as shown. Transistors P43and P44 have gate electrodes coupled to signal SDRV. Thus when SDRV ishigh and, correspondingly state device 72 is providing a logic one to bedriven from the driver 70, two inverter stages later, a logic one willappear on line DISABLE causing P21 to turn on shunting the gate of P22to VSS and thus turning off the turn on charge control circuit 74. TheDISABLE signal also turns off P17.

The turn-on control circuit 74 is also turned off or disabled by thelevel of voltage signal on line IOPROT, rising to above about one voltcausing transistor P19 to turn on coupling CHARGE line to VSS andturning off P22.

The charge current control circuit 74 provides a signal to charge DRVR₋₋GATE line 80a by drawing a current related to the amount of currentdrawn by a representative pair of transistors in the VDACTL controlcircuit (FIG. 9C). This current in turn is related to the current drawnthrough the representative stack in accordance with the value of anexternal resistance provided to set the current through therepresentative stack.

The turn-on charge control circuit 74 provides an output signal coupledto DRVR₋₋ GATE line 80a which feeds the other corresponding circuits asmentioned. The voltage source mode control circuit 73 includes a unitygain amplifier 73a having transistors P3 and P5 with control electrodescoupled respectively to signals VOH and VDACTL. The signal IOPROT iscoupled to the gate electrode of P4 correspondingly the inverting inputof the unity gain amplifier provided by transistors P4, P3, and P5,whereas the signal VOH is coupled to the noninverting input of the unitygain amplifier. Thus signal VOH is used as a reference for the output ofthe unity gain amplifier.

The voltage source mode control circuit 73 further includes transistorP13 having a gate electrode coupled to signal OC₋₋ L provided from theoutput of the I-limit control circuit 77 and having a drain electrodecoupled to DRVR₋₋ GATE line 80a. When the driver 70 is driving a logicone state on output IO, signal OC₋₋ L from the I-limit control circuit77 is initially high causing the transistor P13 to couple the outputfrom the unity gain amplifier (drain of transistor P13) to the lineDRVR₋₋ GATE 80a. Thus, the signal OC₋₋ L from I-limit control circuit 77permits transistor P13 to conduct placing at the output of P13 and onDRVR₋₋ GATE line 80a the current from the output of transistor P3. Thevoltage provided on the DRVR₋₋ GATE line 80a is related to the relativevalues of VOH and IOPROT, and VDACTL which controls the current sourcetransistor P5.

The voltage source mode control circuit 73 further includes transistorsP1, and P6. Transistor P1 has a gate electrode coupled to transistor P2in the turn-off charging current control circuit 76. Transistor P2 alsoprovides for V-source mode operating current from transistor P1 throughcurrent flow in transistors P2 and P32, as well as transistors P7 andP33. The amount of current from P7 and P33 is controlled by the relativevalues of voltage from signals VOH and VDACTL. Transistor P6 has a gateelectrode coupled to signal SDRV₋₋ L and is used to sense the state of alogic signal provided from the output of state device 72.

Accordingly, when state device 72 provides a logic 1 signal, SDRV₋₋ Lwill be at a logic 0 level. This drives transistor P6 to an off statepermitting the voltage source mode control circuit 73 to drive DRVR₋₋GATE line 80a. If however, SDRV₋₋ L is high and thus the state device isdriving a logic zero from the driver 70, then the transistor P6 turnson, coupling the VDD supply voltage to the source connections oftransistors P3 and P4. A voltage level of VDD at the connection of thetransistors P3 and P4 turns off or inhibits both transistors and hencethe voltage source mode control circuit 73, since the difference involtage between VOH and the source of P3 would be insufficient to turnon transistor P3.

Accordingly, signal SDRV₋₋ L controls the state to be driven on theDRVR₋₋ GATE line 80a through transistor P6.

The I-limit circuit 77 (FIG. 9B) includes a unity gain amplifiercomprising transistors P9, P10 and P11 arranged as an amplifier.Transistor P10 is fed by the signal on DRVR₋₋ GATE line 80a at the gateelectrode thereof whereas, transistor P9 has a gate electrode fed bysignal VDRVCTL. I-limit control circuit 77 further includes transistorsP8 and P12, as shown. When the voltage on DRVR₋₋ GATE line 80a is lessthan VDRVCTL, the transistor PS, P10, and P12 conduct placing signalOC₋₋ L in a low state causing the drain electrode of P12 to approachVSS. This signal OC₋₋ L is fed to the voltage source mode controlcircuit 73 disabling the transistor P13 as described, while enablingtransistor P37 in the current source mode control circuit 75. Thus lineOC₋₋ L is used to disable the voltage source mode control circuit 73 andenable the current source mode control circuit 75.

The current source mode control circuit 75 includes transistors P37, P36and P35 arranged to provide a unity gain amplifier type circuit in whichan output voltage at line 80a corresponds to the voltage VDRVCTL. Herethe current source mode control circuit 75 thus provides an outputsignal to drive DRVR₋₋ GATE line 80a to a voltage related to VDRVCTL tomaintain the driver output IO at a constant current in accordance withchanges in load beyond the current limit from devices connected to theoutput line 30a of driver 70.

The output 30a is coupled through a resistor 90a which is part ofreceiver 90 (FIG. 3). Thus the output fed on line 30a produces thesignal IOPROT which is fed to a gate electrode of a transistor P41 inthe voltage limit control circuit 75. The voltage limit control circuit77 further includes transistors P39 and P40 each having gate electrodesfed by signals OC₋₋ L and VOH, respectively. When the voltage IOPROTexceeds VOH and OC₋₋ L is low, the voltage limit control circuit 77stops diverting current through transistor P41 from transistor P39. Acurrent flows through transistor P40 forcing current into the lineDRVR₋₋ GATE line 80a to raise the voltage level of the gate drive to thePMOS drive transistor P45 tending to reduce the output voltage from thedriver device and to return mode of operation to a voltage source mode.

The turn-off charge current control circuit 76 is used to turn off thedriver device 80 and provide a controlled output fall time. The turn-offcharge control circuit is shown to include two stacks of transistors.The first stack comprised of transistors P15, P14, and P16 is fed viasignal SDRV₋₋ L, VOH, and VDACTL, respectively. When SDRV₋₋ L is highand consequently the state device is no longer controlling the driverdevice to drive a logic one, transistor P15 turns on drawing a currentfrom a transistor P2, as shown. A mirror transistor P1 which is part ofthe voltage source mode control circuit 73 souces current which bringsDRVR₋₋ GATE line 80a back to VDD thus turning off the driver device 80and hence providing a controlled fall time at the output of the driver70. The amount of current drawn from P1 and P2 is controlled by therelative values of voltage from signals VOH and VDACTL.

Referring now to FIG. 9C, a control circuit 87 used to provide referencevoltage signal VDACTL is shown to include a terminal RRISE (PAD) towhich is coupled an external resistance 53 (FIG. 2) and an amplifier 87aand to VDD here a compensated amplifier having a low offset voltage. Theexternal resistance 53 is coupled to the non inverting input of theamplifier 87a whereas the inverting input of the amplifier is coupled tothe signal VDRVCTL from FIG. 6. The output of the amplifier is thesignal VDACTL and the signal VDACTL is connected to the control input(gate) of a transistor P11 having a drain connected to the source oftransistor P10 and a source connected to VSS. The gate of transistor P10is coupled to VOH whereas the drain of transistor P10 is coupled to theexternal resistance and the non inverting input of the amplifier 87a.The transistors P11 and P10 are each sized to provide a ratio which isthe same for each VOH and VDACTL pair stack used in the driver 70. Theratio used here is 4.5:1 although other ratios can be used. Thesetransistors P11 and P10 are used as representative devices to provide acompensated signal VDACTL to take into consideration semiconductorprocessing variations as well as changes in electrical characteristicsof the circuits in the IO cell 40 caused by variations in temperature,supply voltage and potentially help compensate for changes incharacteristics caused by aging. The reference voltage signal VDRVCTL isprovided from the output of amplifier 54 (FIG. 6) in accordance with thetransconductance of the transistor P4 (FIG. 6). If the transconductanceof the transistor P4 is low (an indication of a slow process for thebatch of circuits or a high operating temperature) VDRVCTL will be at avoltage further away from VDD. This voltage is provided across RRISEresistor 53 producing a larger current through the stack 88 oftransistors P10 and P11 controlled by VOH and VDACTL.

When VOH and VDACTL are used to control a similar stack of transistorsin the driver or the termination, they will provide a relatively largercurrent from the respective stack.

Correspondingly, if the value of VDRVCTL is closer to VDD an indicationof a fast process or lower operating temperature, the current from thestack 88 and hence corresponding stacks in the driver and terminationwill be lower.

Thus, in the driver, the stack transistors P23 and P24 will provide avariable amount of current from the turn-on charge control circuit 74(FIG. 9) to charge the DRVR₋₋ GATE line at a rate to permit the gate ofthe driver transistor P45 to reach the current limit voltage levelmatching VDRVCTL voltage in a fixed period of time substantiallyindependently of variations in the characteristics of the devices causedby processing, temperature and aging effects.

Referring now to FIG. 10, the overshoot clamp circuit 83 is shown toinclude a positive overshoot clamp circuit 86 and a negative overshootclamp circuit 84. The positive overshoot clamp circuit 86 is used toclamp the output voltage on line IO to a value not greater than themaximum output high voltage VOH plus about 1.0 volts programmed for thedriver circuit 70, whereas, the negative overshoot clamp circuit 84 isused to clamp the voltage on line IO at a value not less than about 0.0volts.

The positive overshoot clamp circuit 86 includes an inverter 86acomprised of transistors P54 and P55 and an output transistor P56. Theinverter 86a is a high threshold inverter, i.e., the size of each of thetransistors is selected to provide a high turn-on threshold fortransitioning from a logic 0 to a logic 1 state. Accordingly, PMOStransistor P54 is preferably approximately ten times larger than theNMOS transistor P55 of the inverter 86a. The inverter is coupled betweenthe IO line 30a and VSS. The input to the inverter is signal VOH. Theoutput of the inverter is taken between transistors P54 and P55 at theirdrains and is coupled to the gate electrode of transistor P56. The drainof electrode P56 is coupled to the bus signal line 30a, whereas thesource of transistor P56 is coupled to the VSS.

Transistor P56 is biased to turn on when the voltage on line IO exceedsthe voltage limit value VOH plus about 1.0 volts. Thus NMOS transistorP56 diverts excess current on the bus line 30a into VSS, clamping thevoltage on line 30a at a maximum voltage of about VOH plus about 1.0volts. When the bus signal voltage falls below the limit value VOH, theinverter 86a turns off NMOS transistor P56 thus inhibiting diversion ofbus signal current into VSS. By using the inverter to control the NMOStransistor P56, the undesirable capacitance commonly associated with adiode type clamp is eliminated. This provides improved, that is, fastersystem operation since there is no recovery switching time associatedwith the transistor as there is with that of a junction diode. Moreover,the excess capacitance loading of the bus line typical of the diodeapproach is reduced by using the transistor 56.

The negative overshoot clamp circuit 84 is shown to include transistorsP28 through P31. Transistors P28 and P29 have gate electrodes fed,respectively, by DRVR₋₋ GATE signal line 80a and signal logic SDRV fromstate device 72 (FIG. 8). The signal DRVR₋₋ GATE is also coupled to thedrain electrode of transistor P31 having a source electrode coupled toline IO. Transistor P28 has a drain electrode coupled to a sourceelectrode of transistor P29 with the drain electrode of transistor P29coupled to the gate of transistor P31, as well as to a resistanceprovided by transistor P30 having its gate electrode coupled to itssource, as shown. Transistor P30 provides a voltage limit connectionbetween transistor P29 and reference potential VSS, analogous to a diodeforward voltage drop above VSS.

Negative overshoot clamp circuit 84 operates as follows: When DRVR₋₋GATE line is driving driver device P45 (FIG. 9), transistor P28 is onsourcing current into P29. Transistor P29 will turn on if SDRV goes lowthat is if the driver 70 discontinues or is not driving a logic onelevel. If a negative voltage is induced on line 30a from the currentflowing in the inherent inductance of the line, then transistor P31 willstay on drawing more current from DRVR₋₋ GATE line 80a. This effectivelydelays the time when the driver device 80 (transistor P45 FIG. 9) turnsoff forcing the driver device 80 (transistor P45) to maintain the line30a above about 0.0 volts. This is the preferable way to limit negativeexcursions because no current is diverted into VSS but rather is stillflowing from VDD until induced current approaches zero.

Referring now to FIG. 11, the speed circuit 82 is shown to include PMOStransistors P17 and P16 having gate electrodes coupled to lines VDRVCTLand PEDALN, respectively. Transistors P17 and P16 are arranged as astack 82a having a source electrode coupled to a drain electrode of asucceeding transistor in the stack 82a. Coupled to the stack 82a is atransistor P15 having a gate electrode fed by signal LZI (FIG. 14).Transistor P17 has its source electrode coupled to voltage VDDTL andtransistor P15 has its drain electrode coupled to line IORES 30a'. LineIORES, as explained earlier in conjunction with FIG. 5, is coupled via aresistance 49b (FIG. 14) to line 30a (IO) (FIG. 9).

The speed circuit 82 further includes a pair of NMOS transistors P12 andP14, with transistor P14 having a drain electrode coupled to line 30a'and a source electrode coupled to the source electrode of transistor P12and transistor P12 has a drain electrode coupled to the drain oftransistor P16 as shown. The gate electrodes of transistors P12 and P14are coupled together and to the drain electrode of transistor P12. Asixth transistor P13 provides a switch to VSS for transistors P12 andP14 and has a drain electrode coupled to the common connection of thesource electrodes of transistors P12 and P14 and, further, has a sourceelectrode coupled to VSS.

The gate electrode of transistor P13 is coupled to terminal LZI theredundant logic output from the receiver circuit 90 (FIG. 14). A currentproportional to the driver current limit controlled by the value ofVDRVCTL flows through the stack 82a (transistors P17 and P16) and inaccordance with the state of the signal on line LZI, current from stack82a will flow either out of line 30a' and thus through line 30a viaresistance 49b or into line 30a (terminal IO) and hence (line 30a')through resistance 49b towards VSS.

When LZI is at a logic 1 state transistor P15 is off and a current fromthe stack 82a flows through transistor P12 to transistor P13. Since thegate of transistor P12 is coupled to the gate of transistor P14 and,further, since the gate is also coupled to the drain of transistor P12,transistor P14 provides a mirror current flow in response to the currentthrough transistor P12. Thus if the transistors are of equal size, asubstantially similar current will flow from terminal IORES throughtransistor P14 into transistor P13 to VSS. Thus, with the state of thelogic signal received on line 30a, at a logic one state current is drawninto the speed circuit 82 causing the voltage on the line 30a to fallbelow that corresponding the level of the received logic high state.

This preconditioning of line 30a aids in reducing the time periodrequired to transition from a logic high state to a logic low state ifthe next state of the bus makes that transition.

For a received logic 0 state, the signal on line LZI turns on transistorP15, thus diverting the current through transistors P17 and P16 throughtransistor P15 and out onto line IORES.

This preconditioning of line 30a aids in reducing the time periodrequired to transition from a logic low state to a logic high state ifthe next state of the bus makes that transition.

Thus, by examining the output from the receiver 90 (FIG. 14), thecurrent from the line IO 30a to IORES 30a' is preconditioned to anassumed opposite next logic state to be driven on the bus line 30a,since the signal LZI corresponds to the true state most recentlyreceived at the receiver 90.

Thus, if the present state received at the receiver is a logic highlevel, then LZI will be at a logic 1 level which will turn offtransistor P15, turn on transistor P13 and divert current throughtransistor P12, producing in response a mirror current throughtransistor P14.

On the other hand, if the data received at the output of the receiver isa logic low level, then LZI will be at a logic 0 level. This will turnon transistor P15, diverting current from transistors P17 and P16through transistor P15 and out to IORES line 30a'.

Since the settling time of a line on a bus is greatest during thosecycles where data is changing from a logic 1 to a logic 0 state or,conversely, from a logic 0 state to a logic i state, the above mentionedpreconditioning of the bus line 30a, (30a') to assume the next logicstate, decreases settling time of the line after the transition since asmaller voltage is necessary to change the state of the line.

A logic signal has a given noise margin or band of voltages about thetransition threshold window which the logic signal must not fall into inorder to differentiate between logic levels. In a cycle where the givendata bit does not change state, the noise margin is very large.Conversely, if the data bit transitions the noise margin limits the bussettling time. At every module bus data is received at each cycleboundary and received state is used to precondition the bus for thefollowing cycle. The preconditioning provided by the speed circuit 82 isprovided on the assumption that a bus transition will occur at the nextstate. Thus the current on the bus is driven towards the assumeddirection, thus moving the bus voltage towards the received thresholdwindow. This has the effect of reducing the noise margin for instanceswhere there is no transition but increases the noise margin(effectively) for instances where there is a transition.

As shown in FIG. 12, the voltage on line IO is not tied to a fixed rigidvoltage. When the first rising clock edge of clock signal CLK provides alogic level 1 at the output of the state device 72 denoted in FIG. 12 atline DRV, this logic state produces the voltage 85a, as illustrated, online IO. Thus voltage on line IO is here shown starting a voltage levelVOL_(c) which is above the voltage level VOL, the minimum voltage for alogic 0 state in the receiver.

At the moment of the transition of driver when the voltage on line IOtransitions quickly up to a voltage VOH, the line IO remains at VOHuntil the receiver and the speed cell 82 switch logic state at nextrising clock edge. The voltage stays at level VOH for one cycle, afterwhich point the voltage drops to a level VOH_(c) which is at a value 85abelow the level VOH, if it is not transitioning low.

As shown in FIG. 12, for the first data cycle, the data changes from thelogic 1 state from the driver to a logic 0 state. The logic 1 state isclocked into the receiver on the next clock cycle CLK2 and the LZIsignal from the receiver is sensed by the speed circuit 82 in the driver70 causing the driver to change the voltage on line IO to drop to thelevel VOL. The voltage stays at VOL for one cycle, at which point thevoltage then rises again to VOL_(c).

During the next clock cycle CLK3, the logic level on the line DRV isagain a logic 0 and, thus, the voltage on line IO stays at VOL_(c). Onthe fourth clock cycle CLK4 the driver transitions to a logic high leveland, thus, the voltage on line IO transitions from VOL_(c) up to VOHremaining at VOH during the first cycle and dropping to VOH_(c) duringthe second cycle.

At each one of the points 85a, 85b the speed circuit 84 preconditionsthe bus for the next logic state which is assumed to be a logic lowlevel, thus dropping the voltage on the bus from the level VOH toVOH_(c) or raising the voltage from VOL to VOL_(c).

This preconditioning of the bus 30 provides an improvement in the speedon the bus 30. The improvement in speed is related to the value at whichVOL_(c) and VOH_(c) are set. These values, in turn, are related to thevalue of the driver current limit controlled by the voltage VDRVCTL.This current limit is proportional to voltage reference VIH divided byexternal resistance RTERM (FIG. 6).

Referring now to FIG. 13, the logic state device 72 is shown to includea master latch 72a and a slave latch 72b. The state device 72 furtherincludes an inverter 72c which is fed via the output signal SDRV fromslave latch 72b. The inverter 72c comprised of a PMOS and NMOStransistor provides signal TENB which is used to disable the terminationassociated with the particular driver in the IO cell when the output ofa state device 72 is in a state that will cause the associated driver ofthe particular IO cell to drive its own IO line. Here the driving stateof the cell is a logic 1 level. Thus when signal SDRV is a logic 1level, transistor P61 conducts causing line TENB to go to a logic0,disabling termination.

The master latch 72a receives data at terminal DATA and a clock signalat terminal CLK. Assuming that enabling signal at terminal EN is in astate which enables one data , the data is latched in master latch 72aand is presented at the output at M DATA and is latched by the slavelatch 72b at a next falling clock edge. In general, any conventionalmaster-slave flipflop would be suitable for logic state device 72.

Referring now to FIG. 14A, the receiver 90 (FIG. 3) is shown to includea latch 90a coupled directly to the bus line IO of the bus 30 (FIG. 1).The latch is also fed via a system or bus clock signal CP used to clockin data to the latch 90a and a reference voltage REFSINK from which areference threshold voltage level is provided. The outputs from latch90a are a latched, unamplified data bit from line IO and a latchedreceiver reference voltage derived from the signal REFSINK. Thesesignals are amplified by an amplifier type of device 90b and are thenfed to the second latch 90c. At the outputs of the second latch 90c arethus provided the signals ZI and LZI.

The receiver 90 is a master-latch/amplifier/slavelatch arrangement. Thisarrangement has the unamplified signal from the bus latched in the latch90a prior to being fed to a state determining circuit or an amplifier.This arrangement of latching before amplifying eliminates the setup timeassociated with prior receiver arrangements where an amplified bussignal is provided before entry into a master-slave latch. Afterlatching of the unamplified bus signal, the signal is amplified andresolved to provide a determination of the represented logic state inaccordance with a receiver reference voltage transition thresholdprovided from REFSINK. The resolved and amplified logic state is thenlatched into the slave latch 90c, providing in response the outputs ZIand LZI. Referring now to FIG. 14B, the receiver 90 is shown to includean event sequencer circuit 92 which is fed via an inverter 91, anopposite phase CLKL of the external clock signal CP from the systememploying the receiver 90. The clock phase CLKL is provided from theoutput of inverter 91 and is fed to the event sequencer circuit 92. Theevent sequencer circuit 92 provides in response to the clock signalCLKL, clock phases RC1H, RC2L, RC3H, RC4L, RC5H and RC6L, as shown. Eachof the signals RC1H through RC6L are selectively delayed alternatingphased clock signals and are used to control the clocking of events ineach of the circuits to be described.

Here an exemplary timing diagram showing the relationship between theaforementioned clock signals is illustrated in conjunction with FIG.15A. Each succeeding one of the clock phase signals CLKL, RC1L etc. issuccessively delayed a predetermined amount in accordance with the sizeof the inverter provided to generate the respective signal. These delaysare used in the receiver 90 in order to control timing of events in thedevice.

The event sequencer circuit 92, fed by clock signal CLK₋₋ L, providessignals RC1H through RC6L to a data resolution circuit 94, a powercontrol circuit 95 and an output latch and buffer circuit 96 as needed.The data resolution circuit 94 is also fed via signal IO (FIGS. 3 and14A) and via a switching threshold reference signal RCVREF to provide inresponse thereto differential output logic states on BIBUS and BIREF, asshown. The data resolution circuit 94 is further fed switched power andreference signals SWTCH₋₋ VDD and SWTCH₋₋ VSS, as well as clock phasesignals CLKL, RC2L, RC3H and RC5H, as also shown.

The data resolution circuit 94 as will be described in conjunction withFIG. 16, is controlled by clock phases CLK₋₋ L, RC2L, and RC5H and RC6L.The data resolution circuit 94 latches data directly from the bus lineIO and determines or resolves the logic state of the latched dataproviding differential logic output signals on lines BIBUS and BIREF.

The output latch and buffer 96 receives the logic state signals on lineBIBUS and BIREF and provides in response thereto an output on signal ZIwhich is used by the remainder of the application specific integratedcircuit (not shown), as generally described in conjunction with FIG. 2.The output latch and buffer circuit 96 further provides signal LZI whichis here dedicated to use by the speed circuit cell (FIGS. 8 and 11), asgenerally described above. The output latch and buffer circuit 96 iscontrolled via clock phase signals RC2L and RC3H, as also shown.

The receiver is here a low setup time, low propagation delay device;i.e., the time required for the signal on line IO to propagate throughthe circuit and be valid after the reception of the clocking edge ofsignal CP is relatively small. Here the setup time is approximatelyminus 50 pico seconds or essentially there is no setup time requirement.Here also the hold time is about 300 picoseconds. The receiver 90 isfurther characterized as having a relatively low propagation delay;i.e., the delay through the circuit 90 between the signal on IO relativeto the clock phase CP to the output zI. Here the propagation delay is onthe order of less than 1.5 nanoseconds.

Referring now to FIG. 16, the data resolution circuit 94 (FIG. 14B) isshown to include an input latch comprised of circuits 102a and 102b, asshown, which are coupled, respectively, between line IO and line RCVREFand a resolver circuit 104. Input latches 102a, 102b are used to latch avoltage level associated with signals on line IO and line RCVREF at aselected time period in accordance with timing signals provided viaclock signal CLKL.

The latches 102a and 102b are thus used to isolate the signals IO andRCVREF, as captured by the data resolution circuit 94, from theremainder of the receiver 90 during a data resolution phase of thereceiving cycle for receiver 90. In particular, input latches 102a, 102bare used to isolate the resolver circuit 104 from the remainder of thereceiver and the system incorporating the receiver.

In some applications of the receiver 90 it is preferable to bypass theline RCVREF to VDD and VSS with capacitors 111a, 111b, as shown in FIG.18 provided from preferably PMOS or NMOS transistors having drain andsource electrodes coupled together. Other control and reference signalsmay also benefit from this dual coupling arrangement in order toadequately compensated for common mode type noise on signal lines.

The resolver circuit 104 can be either of several embodiments. Inparticular, resolver circuit 104 can be a pair of cross coupledinverters as will be further described in conjunction with FIG. 17 or,alternatively, can be a differential amplifier. Other arrangements forresolver circuit 104 may alternatively be used but here the preferredarrangement is that as will be described in conjunction with FIG. 17.

The data resolution circuit 94 further includes precharge circuits 106aand 106b which have an output coupled to line REFSINK and which have anoutput also coupled to inputs 104a, 104b of resolver circuit 104.Precharge circuits 106a and 106b are used to precharge the resolvercircuit particularly as will be described in conjunction with FIG. 17 toa known charge state prior to latching of new data via input latches102a, 102b. Thus, the precharge of circuit 104 is used to provide theresolver circuit 104 to a known reference and, furthermore, is used todischarge excess charge provided in the resolver circuit due to aprevious received state.

The data resolution circuit 94 further includes isolation latches 108aand 108b which are disposed between corresponding input latches 102a,102b and respective terminals IO and RCVREF. Isolation latches 108a and108b are used to provide additional isolation for the data resolutioncircuit 94 particularly after CLKL transitions to open up latches 102aand 102b, as will be further described below.

The outputs from resolver circuit 104 are fed into output latches 109aand 109b which are used to isolate the outputs of resolver circuit 104from the remainder of the receiver 90. In particular, output latches109a and 109b are used to latch data as resolved from resolver circuit104 and hold said data while a new receiver cycle is initiated inresolver circuit 104.

Referring now to FIG. 17, a preferred implementation for resolvercircuit 104 is here shown to include a pair of cross coupled inverters104a' and 104b', as shown. Here inverter 104a' includes a PMOStransistor P3 coupled to an NMOS transistor P28. The drain electrode ofPMOS transistor P3 is coupled to the drain electrode of transistor P28and is also coupled to input 104a. The input 104a is also coupled togate electrodes of transistors P15 and P44 of the second inverter 104b'.The second inverter 104b' further has the drain electrode of transistorP15 coupled to the drain electrode of transistor P44 and is also coupledto a second input 104b to the resolver 104. Similarly, the input 104b iscoupled to the gate electrodes of the transistors P3 and P28 of thefirst inverter 104a'.

Signals SWTCH₋₋ VDD and SWTCH₋₋ VSS are also coupled to hererespectively a common connection of the source electrodes of transistorsP3 and P15 and a common connection of the source electrodes oftransistors P28 and P44. Power and reference thus are selectivelyprovided to the resolver circuit 104 in a time sequence which permitsthe resolver circuit 104 to make a quick determination as to therelative logic state represented by a signals on line 104a relative to areference on line 104b, as will be described.

Referring now to FIG. 18, an exemplary illustration of a preferredembodiment of the receiver 90 is shown to include the aforementionedevent sequencer circuit 92, data resolution circuit 94, power controlcircuit 95, and output buffer and latch circuit 96. Here the eventsequencer circuit 92 is shown to include a plurality of here six stagesof inverters coupled together, as shown, with the outputs of here thesecond and succeeding ones of said inverters providing theaforementioned clock phase signals RC1H through RC6L, respectively. Thesize of each of the transistors providing the CMOS inverter arrangementsis chosen in order to provide corresponding successively increasingdelays with respect to clock signal CLKL. An illustrative example of thesuccessive delays between each of the clock phases is illustrated inconjunction with FIG. 15A, as mentioned above. FIG. 15A will be referredto as necessary to explain further operation of the receiver 90.Moreover, FIG. 15B shows a typical sequence of timing in the receiver 90for each of the stages of operation of the receiver.

The signals RC1H through RC6L are coupled or fed as necessary to theremaining circuits in the receiver 90. Thus, the data resolution circuit94 which is shown to include the input latch 102a and 102b here eachprovided as an NMOS transistor P24 and P23 coupled in a pass transistorconfiguration between the respective inputs IO, RCVREF and inputs 104aand 104b to resolver 104. That is the transistors P24 and P23 allowrespective signals from lines IO and RCVREF to pass through as long asthe control electrode or gates are high. When the gates are low, thetransistors are turned off and any charge provided between respectivesource electrodes and inputs 104a and 104b is isolated from the inputsof the receiver 90. As shown in FIG. 15B, when CLKL is low thetransistors P24 and P23 isolate the state in the resolver circuit 94.

The data resolution circuit further has the precharge circuits 106a and106b here provided as stacks of NMOS transistors P51, P52 and P29 andP30, respectively, having outputs coupled to the inputs 104a and 104b,respectively, of data resolver 104. Also shown as coupled in a passtransistor configuration are transistors P25 and P22 which respectivelyprovide isolation latches 108a and 108b.

The outputs IBUS and IREF from the resolver 104 are fed to outputisolation buffers 109a and 109b, respectively, as shown, here providedby a stack of transistors P4, PS, P31 and P32 for latch 109a and a stackof transistors P7, P6, P33 and P34 for latch 109b. The outputs of theseoutput isolation buffers are fed to output latch 96 which is used tolatch the relative differential state provided on lines BIBUS and BIREFand to maintain that state after removal of signals from the buffers109a and 109b.

Power control circuit 95 is also shown to include a stack of seriescoupled transistors P8, P35, P65, P64, P37 and P36, as shown, andcontrolled via clock signals CLKL, RC5H and RC1H as also shown. Ingeneral, in accordance with the timing sequence provided by theaforementioned clock signals, the switch SWTCH VDD and SWTCH₋₋ VSSsignals are sequentially applied to the resolver circuit 104 after therising CP clock edge.

With particular reference to FIG. 15B, the receiver 90 operates asfollows:

A signal is provided from the bus onto bus line 30a to terminal IO asgenerally mentioned above. When the signal is fed to terminal IO to thereceiver 90, latch transistor 102a is in a pass mode; i.e., the clockphase CLKL is high causing the PMOS transistor P24 to conduct the signalthrough the transistor to the input terminal 104a of resolver 104.Similarly, the pass transistor P25 is also "on" or high since the signalRC6L is also in a high state during this initial period.Correspondingly, latch transistor 102b and latch transistor 108B arealso in a pass mode allowing a voltage corresponding to signal RCVREF topass through the aforementioned transistors to terminal 104b.Thereafter, when clock signal CP (FIG.15B) transitions high, i.e., froma logic 0 to a logic 1 state, the data on lines IO is valid and thusintended to be clocked into the receiver 90. Correspondingly, signalCLKL is clocked low causing the transistors P24 and P23, i.e., latches102a and 102b, to open holding or isolating charge on the inputs 104aand 104b of the resolver 104 and isolating the resolver 104 from theinput terminal IO and input terminal RCVREF.

At the initial portion of the receive cycle, nodes SWTCH₋₋ VDD andSWTCH₋₋ VSS are at an initial state of 0.6 volts as provided from theprecharge circuits 95, 106a and 106b during the previous cycle and aswill be discussed shortly. Assuming that a logic 1 is clocked in on lineIO, then the signal on the input 104a is a voltage corresponding to avoltage level above a predetermined amount above the reference voltageRCVREF.

In the illustrative example, the reference voltage is 0.6 +/-0.025volts. Thus a voltage greater than 0.625 volts is required todifferentiate a logic 1 from a logic 0 state and a voltage of less than0.575 volts is required to differentiate a logic zero state from a logicone state. The band of voltages +/-0.025 volts are the result of circuitand layout capacitance coupling imbalances. Preferably, there is a smallamount signal amplitude required for fast data resolution and to avoidresolver metastability.

Thus here when receiving a logic one state the voltage on line 104a isillustratively between 0.625 volts and 1.5 volts. The voltage at thesource electrode of transistor P3, as well as the source electrode oftransistor P28, is 0.6 volts based upon the precharge condition providedas mentioned above. When the voltage from the data latch 102a and thatfrom latch 102b are each fed to the resolver 104, each of the inverters104a' and 104b' are preferentially disposed to transition in oppositedirections in accordance with the voltage provided at terminals 104a and104b.

At the source electrode at transistor P3 (coupled to SWTCH₋₋ VDD), avoltage of 0.6 volts (rising) is provided, whereas, at the sourceelectrode of transistor P28 (coupled to SWTCH₋₋ VSS) a voltage of 0.6volts (falling) is provided. When CLKL transitions low, the transistorP8 in power control circuit 95 is driven on causing the signal SWTCH₋₋VDD to approach voltage VDD, the supply voltage, as shown in FIG. 15B.Thus the supply voltage is initially coupled to the source of transistorP3 and the source of transistor P15. Correspondingly, transistors P35and P37 in the power control circuit are driven off. A short time delaythereafter, signal RC1H is asserted causing transistor P36 in powercontrol circuit 95 to transition on coupling signal SWTCH₋₋ VSS to VSSor a reference voltage as shown in FIG. 15B.

After this initial sequence, supply voltage SWTCH₋₋ VDD and referencevoltage SWTCH₋₋ VSS are coupled to the resolver 104. In responsethereto, the gate electrodes of transistors P3 and P28 are at areference of 0.6 volts, whereas, the gate electrodes of transistors P15and P44 are at a voltage corresponding to a logic 1 from the latcheddata provided from terminal IO (here 0.625 volts up to 1.5 volts). Atthis point transistor P15 and transistor P44 each have a gate electrodecoupled to terminal 104a which is at a high logic level in comparison tothe voltage at RCVREF and transistor P3 and transistor P28 each have agate electrode coupled to terminal 104b which is at RCVREF.

The voltage on line RCVREF causes transistor P3 to turn on which furtherenforces the logic 1 state at the node 104a and provides a correspondinglogic 1 output on signal line IBUS. This value of voltage on the gate oftransistor P44 will keep transistor P44 on causing a reinforcement of alogic 0 state on line 104b and output line IREF, thereby reinforcing thestate on the gate of transistor P3. Similarly, the voltage on the gateof transistor P15 will be relatively high causing transistor P15 to stayoff and the relatively low voltage on transistor P28 will keeptransistor P28 off. Thus a logic 1 level will be provided on IBUS and alogic 0 level will be provided on IREF.

A similar arrangement is provided for a logic 0 latched in at node 104a.At this point transistor P15 and transistor P44 each have a gateelectrode coupled to terminal 104a which is at a low logic level incomparison to the voltage at RCVREF, and transistor P3 and transistorP28 each have a gate electrode coupled to terminal 104b which is atRCVREF.

The voltage on line 104b causes transistor P3 to turn off which furtherenforces the logic 0 state at the node 104a and provides a correspondinglogic 0 output on signal line IBUS. This value of voltage on the gate oftransistor P15 will keep transistor P15 on causing a reinforcement of alogic 1 state on line 104b and output line IREF, thereby reinforcing thestate on the gate of transistor P28. Similarly, the voltage on the gateof transistor P44 will be relatively low causing transistor P44 to stayoff and the relatively high voltage on transistor P3 will keeptransistor P3 off. Thus a logic 0 level will be provided on IBUS and alogic 1 level will be provided on IREF at the transition as indicated inFIG. 15B.

In any event, after the logic state is resolved by the resolver 104, thesignals on lines IBUS and IREF are fed to buffers 109a and 109b,respectively. During the period of time when RC2L is low and RC3H ishigh as shown in FIG. 15B, either transistor P5 or P32 in buffer 109a ortransistor P34 or P7 in driver 109b will be turned on to couple therespective logic level to outputs BIBUS and BIREF, respectively.

Signals on BIBUS and BIREF are thus buffered signals provided by outputlatch buffers 109a and 109b which are interposed to isolate the resolver104 from the output stage or latch 96 of the receiver 90. These signalsare fed into the output latch 96.

The output latch 96 includes a first inverter 97a comprised oftransistors P11 and P40, as shown. The signal BIREF is fed to the gateelectrodes of transistors P11 and P40 providing an output correspondingto the opposite state from BIREF. This output is likewise coupled tosignal BIBUS which reinforces the output of the inverter. With thisarrangement, a single inverter stage 97a (transistors P11 and P40) isused to single-endedly latch the differential resolver outputs BIBUS andBIREF, while still maintaining complete symmetry through the dataresolver 104 and the data resolution circuit 94.

That is, symmetry is maintained between the receiver node IO and thereference node RCVREF. This symmetry is desirable to insure proper andfast operation of the data resolver 104 by maintaining a charge balanceon the pair of nodes 104a and 104b which is upset only by the state of asignal coupled to the node 104a from the terminal IO.

Returning to the output latch 96, the output from inverter 97a is fedthrough a second inverter stage 97b comprised of transistors P19 and P48to provide output zI, a true output corresponding to the value receivedat terminal IO, as shown in FIG. 15B. Similarly, this output is also fedto a third inverter stage 97c comprised of transistors P1, P10, P39 andP26. This third inverter stage 97c is controlled via transistors P10 andP39 when clock signal RC3H is low and RC2L is high. The gates totransistors P1 and P26 are fed via the output of the first inverter pairP11 and P40 and the drain electrode of transistor P10 is likewisecoupled to the gate electrodes P11 and P40. Thus, this switched inverteris used to maintain the logic state of the inverter P11 and P40, hence,inverter P19 and P48 after signals BIBUS and BIREF disappear to enable adevice (not shown) coupled to terminal ZI to clock in the state of asignal received at terminal ZI. A fourth inverter stage 97d is providedvia transistors P38 and P9 coupled to the first inverter output P11 andP40. This inverter 97d is used to provide an additional high assertioncopy (LZI) of the receiver output (ZI) for the speed cell circuit 82(FIG. 8 and 11). This dedicated signal for the speed circuit ispreferred so that the signal is not affected by the loading effectsassociated with the application specific circuit (ASIC) which receivesdata from terminal ZI.

Referring now to FIG. 19, a low offset voltage, high bandwidthoperational amplifier 120 suitable for use as the difference amplifierin the control circuit of FIG. 6, 7 and 9C for example is shown toinclude a first or primary amplifier 122 having non-inverting andinverting terminals 122a, 122b, an enable terminal 122c for testingpurposes, output terminal 122d and a bias input terminal 122e. Theinputs of the amplifier 122 provide the inverting and non-invertinginputs to the amplifier 120. The amplifier 120 further includes twoadditional amplifiers 124 and 126 each having like inputs and an outputas the aforementioned amplifier 122.

Amplifier 124 acts an a replicate amplifier, and has its inverting andnon-inverting inputs fed by a reference voltage REF. The reference inputsignal is here either the INV or NINV input of amplifier 120. Theamplifier 124 produces an output voltage in response to the zerodifferential input voltage from the connection of the inputs to eachother. This output voltage which is the offset voltage of the amplifier124 and which is comparable in magnitude to that of the amplifier 122 isfed to the inverting input of the error amplifier 126. Coincidentally,the output of amplifier 122 is fed to the non-inverting input ofamplifier 126. In response, a signal CAL is provided at the output ofthe amplifier 126 which is used to adjust the bias on a transistor inamplifiers 122 and 124 to compensate for the offset voltage error in theamplifier 124 and amplifier 122. That is if the offset voltage of theamplifier 124 is relatively high, the signal CAL is correspondingly highbiasing the replicate amplifier to produce the same output voltage asthe primary amplifier 122. The signal CAL also biases the amplifier 122at a higher level to thus compensate for the higher offset voltage.

Each of the amplifiers 122, 124, and 126 are preferably identical or atleast designed to provide substantially identical offset voltagecharacteristics. It is thus preferred therefore that the amplifiers arefabricated on the same semiconductor substrate.

Referring now to FIG. 20, a preferred implementation of each of theamplifiers 122, 124, 126 of FIG. 19 is shown to include a differentialamplifier stage 132 having input transistors P4 and P5 each having gatesproviding respectively, the inverting and non-inverting inputs to theamplifier 122. The transistors P4 and P5 have drain electrodes coupledto respective drain electrodes of PMOS transistors P6 and P7, as shown.The gates of transistors P6 and P7 are likewise coupled to the drainsthereof, as shown. With the voltage on the gate of transistor P1 lowtransistors P1 and P2 generate a current flow and mirror voltage for atransistor P3 which is coupled as a current source for transistors P4and P5. That is, transistor P3 provides a constant current source forthe differential stage 132 and transistors P6 and P7 are here connectedas current mirror input structures having gate electrodes coupled to thedrain electrode. These transistors P6 and P7 are used to provide gateelectrode voltage and hence operating drain current control to P8 andP10.

The amplifier 122 further includes a pair of translation and mirrortransistors P8 and P9. Transistor P8 also has its drain electrodecoupled to a gate electrode of a transistor P11 having a drain electrodecoupled to the output terminal 122d. Transistors P10 and P11 are used toproduce mirror currents into or out of output terminal 122d, in responseto current changes in the differential amplifier stage 132 and as willbe described in response to changes in the level of the signal onterminal BIAS. These mirror currents provided from transistors P11 andP10 into or from the output terminal OUTPUT 122 thus provide theamplifier output current.

The signal on terminal BIAS is fed to a gate electrode of a transistorP12, as shown. The drain of transistor P12 is coupled to a commonconnection of the drain of transistor P5 the drain of transistor P7 andgate of transistor P10. A current flow through transistor P12 inaccordance with the level of the voltage on the gate of transistor P12produces in response a corresponding current flow through transistor P7.Since the gates of transistors P7 and P10 are tied together, a mirrorcurrent flow in accordance with the change in current flow through thetransistor P12 is provided through transistor P10 to the output 122d.This compensates for a naturally negative offset voltage characteristicof the amplifier. For a naturally positive offset voltagecharacteristic, a similar arrangement can be used to change current intransistors P4 and P6 and thus vary current flow through transistor P11.

Having described preferred embodiments of the invention, it will nowbecome apparent to those of skill in the art that other embodimentsincorporating its concepts may be provided. It is felt therefore thatthis invention should not be limited to the disclosed embodiments butrather should be limited only by the spirit and scope of the appendedclaims.

What is claimed is:
 1. A bus interface circuit, comprising: a pluralityof I/O cells, each cell comprising:termination means, responsive to avoltage reference signal representative of a desired impedance value,for providing the desired termination impedance value for the cell; andsaid bus interface circuit coupled to said termination means comprises:means, coupled to a resistance, for providing said voltage referencesignal to said termination means of each of said I/O cells in responsiveto said resistance.
 2. The circuit of claim 1 wherein said terminationmeans is a first termination means and said voltage reference is a firstvoltage reference and wherein said means for providing the first voltagereference signal comprises:second termination means, substantiallyidentical to said first termination means in each of said I/O cells andresponsive to a second voltage reference signal corresponding to apredetermined system voltage reference supply, for providing the firstvoltage reference signal, by providing a termination impedance at theoutput of said second termination means corresponding to saidresistance.
 3. The circuit of claim 2 wherein said second terminationmeans comprises:means, responsive to control signals, for providing afirst portion of the desired impedance; and a fixed impedance connectedto said means for providing a first portion of the desired impedance. 4.The circuit of claim 3 wherein said first voltage reference signal isthe same signal fed to the first termination means of each of said I/Ocells.
 5. The circuit of claim 4 wherein said circuit is fabricated asan integrated circuit and said resistance is a single resistor disposedexternal to the integrated circuit.
 6. The circuit of claim 1 whereinsaid termination means comprises:means, responsive to control signals,for providing a first portion of the desired impedance; and a fixedimpedance connected to said means for providing a first portion of thedesired impedance.
 7. The circuit of claim 6 wherein said voltagereference signal is the same signal fed to the termination means of eachof said I/O cells.
 8. The circuit of claim 6 wherein said means forproviding said first portion of said impedance comprises:means,responsive to said voltage reference, for providing a plurality ofsignals representative of the first impedance; and a plurality of activedevices each having a control electrode, and a pair of outputelectrodes, with the control electrode of each device being fed by acorresponding of said plurality of signals.
 9. The circuit of claim 8wherein each I/O cell of said circuit further comprises:a driver havingan output terminal; and a receiver having an input terminal, with saiddriver output terminal and said receiver input terminal being connectedwith an output of said termination means, to provide the terminationresistance at an output of the cell.
 10. A bus interface circuit,comprising: a plurality of input/output bus cells, each cellcomprising:a first resistance termination circuit responsive to a firstvoltage reference signal representative of a desired resistance valuefor the circuit to provide the desired termination impedance value forthe cell, and wherein said bus interface circuit comprises: means,coupled to said first resistance termination circuit, for providing saidfirst voltage reference signal to said first termination circuit of eachof said I/O cells in responsive to a resistance said means including: asecond resistance termination circuit, substantially identical to saidresistance termination circuit in each of said I/O cells and responsiveto a second voltage reference signal corresponding to a predeterminedsystem voltage reference supply to provide the first voltage referencesignal by providing a termination resistance at the output thereofcorresponding to said resistance.
 11. The circuit of claim 10 whereinsaid second resistance termination circuit comprises:means, responsiveto control signals, for providing a first portion of the desiredimpedance; and a fixed impedance connected to said means for providing afirst portion of the desired impedance.
 12. The circuit of claim 11wherein said first voltage reference signal is the same signal fed tothe first resistance termination circuit of each of said I/O cells. 13.The circuit of claim 12 wherein said circuit is fabricated as anintegrated circuit and said resistance is a single resistor disposedexternal to the integrated circuit.
 14. The circuit of claim 13 whereinsaid first voltage reference signal is the same signal fed to the firsttermination means of each of said I/O cells.
 15. The circuit of claim 14wherein said means for providing said first portion of said impedancecomprises:means, responsive to said voltage reference, for providing aplurality of signals representative of the first impedance; and aplurality of active devices each having a control electrode, and a pairof output electrodes, with the control electrode of each device beingfed by a corresponding of said plurality of signals.
 16. The circuit ofclaim 15 wherein each I/O cell of said circuit further comprises:adriver having an output terminal; and a receiver having an inputterminal, with said driver output terminal and said receiver inputterminal being connected at with an output of said resistancetermination circuit, to provide the termination resistance at an outputof the cell.